2. strip tunable stop-band functions using gyromagnetic absorp-
tion and requiring low dc magnetic field are reported. This
article is structured as follows: in Sec. II, the multilayered
material is briefly described and the propagation structure
established. Section III is devoted to the electromagnetic
analysis of the microstrip line containing the ferromagnetic
composite. Finally, Sec. IV presents experimental results ob-
tained with tunable microwave devices.
II. DESCRIPTION OF THE MICROSTRIP STRUCTURE
A. Magnetic composite material
The composite material we have integrated in microstrip
circuitry consists of alternated ferromagnetic and insulating
thin films ͑Fig. 1͒. This material characterized by its relative
effective permittivity eff and permeability eff , exhibits in-
trinsic impedance Z ͑2͒ larger than unity over a wide range
of frequencies for one polarization of the incident wave,7
whereas both matrix and inclusions have an intrinsic imped-
ance lower than unity. Moreover, when applying an external
dc magnetic field, the values of Z can change with the mag-
netic field dependent permeability i(H0) of ferromagnetic
thin films. This composite is called Laminated Insulator Fer-
romagnetic on the Edge ͑LIFE͒ material.7
Zϭͱ0eff
0eff
. ͑2͒
It is made of amorphous CoNbZr materials deposited
onto 12.7 m thick polyimide ͑Kapton͒ by magnetron sput-
tering. The deposition technique for the CoNbZr thin films
has already been thoroughly discussed in Ref. 8. The thick-
ness of the ferromagnetic layer is 0.43 m. As a conse-
quence, skin effect has no significant influence in the centi-
meter wave range. The deposited films have uniaxial in-
plane anisotropy. The uniform orientation of the easy axis on
most of the film surface is due to the influence of the static
magnetic field of the planar magnetron. Thus, CoNbZr films
are anisotropic materials and their permeability is a tensorial
quantity. However, the very small thickness of the film
makes the off-diagonal component of the permeability tensor
of LIFE composite negligible when the microwave magnetic
field is parallel to the plane of the layer. So the magnetic
properties of LIFE material can be described by a scalar
effective permeability eff . Saturation magnetization of the
CoNbZr films is of about 11.3 kG and anisotropy field is of
33 Oe. Thus, the gyromagnetic resonance frequency fr is
about 2 GHz in a completely demagnetized state ͑H0ϭ0 Oe͒.
This result will be experimentally verified and discussed fur-
ther on.
A LIFE composite is manufactured from 37 CoNbZr/
kapton sheets glued together with an epoxy resin. Its relative
effective permittivity eff and permeability eff depend on
the polarization of the incident wave. When the LIFE mate-
rial is lighted on its edge by a microwave radiation polarized
with the electric field perpendicular to the laminations and
the magnetic field perpendicular to the easy axis of the fer-
romagnetic layers ͑Fig. 1͒, eff is low and eff is high. Then,
the intrinsic impedance Z of the composite is larger than
unity and can be successfully exploited for microwave appli-
cations. But, when the electric field is parallel to the lamina-
tion plane, the LIFE material behaves as a good conductor
and exhibits very large conduction losses at microwave fre-
quencies. In the first configuration, eff and eff can be easily
calculated using Wiener’s law9
as
effϭ
m
1Ϫq
and effϭq͑i
Ϫ1͒ϩ1, ͑3͒
where m is the relative permittivity of kapton and glue
͑mϭ2.2͒, i is the relative permeability of CoNbZr layer
͑the static value of i is 340 in a completely demagnetized
state͒, and q the volumic fraction in ferromagnetic material
͑qϭ2.51 %͒. The frequency-dependent i is obtained by us-
ing the Bloch–Bloembergen model.10
Broad-band measure-
ments of eff and eff on a rectangular LIFE sample using an
asymmetrical strip transmission line11
have permitted to vali-
date the calculated values for eff and eff . In Fig. 2, theo-
retical and experimental eff data as function of frequency
are compared in a completely demagnetized state. Calculated
and measured values for eff and eff are in good agreement
on the whole frequency band. The averaged resonance line-
width (⌬H) of the LIFE composite is determined from this
FIG. 1. Description of the LIFE material polarized on the edge.
FIG. 2. Measured effϭeffЈ ϪjeffЉ and effϭeffЈ ϪjeffЉ data vs frequency
for a demagnetized rectangular LIFE sample. Comparison of the measured
and calculated effϭeffЈ ϪjeffЉ data.
5450 J. Appl. Phys., Vol. 91, No. 8, 15 April 2002 Salahun, Que´ffe´lec, and Tanne´
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3. measurement: ⌬Hϭ390 Oe; this value corresponds to a
damping parameter of ␣ϭ0.057. The quality factor of the
material Q0 is calculated using
Q0ϭ
2..fr
␥.⌬H
ϭ
1
2.␣
. ͑4͒
Then, for the LIFE composite, Q0 is equal to 8.78. This
quantity is an important parameter for the loaded quality
factor of the resonator based on the gyromagnetic
phenomenon.12
To improve the quality factor of the material,
ferromagnetic fraction of the composite can be decreased. It
leads to a sharper gyromagnetic resonance. We can notice in
Fig. 3 that eff is clearly disturbed by a dc magnetic field.
During the experimental process, the dc magnetic field is
applied along the wave propagation direction which corre-
sponds to the easy axis direction of the ferromagnetic layers.
The gyromagnetic resonance frequency shifts from 1.8 up to
6 GHz with an increase of the external field H0 from 0 to
350 Oe.
This material is characterized by good static magnetic
properties: a low coercivity ͑HCϭ1.2 Oe͒, a high remanence
͑4Mrϭ4435 G͒ and a square hysteresis loop ͑Fig. 4͒; all
these properties are, indeed, required for microwave
applications.13
The magnetization measurement of the mate-
rial shows that the hysteresis loop corresponds well to a
highly anisotropic system.
In the next paragraph, we will show how to exploit these
properties in a microstrip line to realize magnetically tunable
stop-band functions.
B. Insertion of the composite in a microstrip line
The main problem in the insertion of a LIFE sample in a
microstrip line is to find a configuration for which the mate-
rial is lighted by a correctly polarized wave on the edge in
order to achieve a high intrinsic impedance and low conduc-
tion losses in the structure. The microwave magnetic field hrf
must be parallel to the ferromagnetic layers and the electric
field erf perpendicular to them. Moreover, in order to inten-
sify the gyromagnetic effect, hrf must be perpendicular to the
easy axis of ferromagnetic films, e.g., perpendicular to mag-
netic moments.
The fundamental mode in the microstrip line is a quasi-
transverse electromagnetic ͑TEM͒ mode. The field pattern of
this mode led us to study and compare two locations for the
LIFE material in the microstrip structure. The first one con-
sisted in positioning a rectangular LIFE sample on the strip
conductor of the line. This configuration results in a low
interaction between the wave and the material since no evi-
dent gyromagnetic absorption of the microwave is observed
in the measurement of the absorbed energy depicted in Fig. 5
͑dashed line͒.
The other configuration we tested consisted in inserting a
rectangular piece of LIFE composite into a dielectric sub-
strate between the strip and the ground plane of the line as
sketched in Fig. 6. A low permittivity substrate ͑rϭ1.07͒
composed of a compressible foam14
was employed to con-
centrate the main part of the microwave power in the LIFE
composite. The substrate thickness was of 635 m ͑standard
value͒. Access lines had been designed for a characteristic
impedance of 50 ⍀ to match those of the cables and of the
network analyzer system used to measure the scattering pa-
rameter ͑S parameters͒. The center conductor width calcu-
lated by MDS software15
was 3.3 mm when the substrate was
the foam material. In this configuration, when H0ϭ0 Oe the
transmission response of the microstrip line reveals a peak
absorption in the vicinity of the gyromagnetic resonance fre-
quency ͑Fig. 5͒. It can be noticed that the measured fre-
quency where maximal absorption occurs ͑3.5 GHz͒ is not
the same as the calculated one ͑frϭ2 GHz͒. The main reason
FIG. 3. Microwave permeability of a LIFE sample under various static
magnetic fields.
FIG. 4. Normalized hysteresis loops measured at room temperature with the
field applied parallel ͑plain line͒ and perpendicular ͑dotted line͒ to the easy
axis.
FIG. 5. Measurement of the absorbed energy in the microstrip line as func-
tion of frequency for different positions of the LIFE sample.
5451J. Appl. Phys., Vol. 91, No. 8, 15 April 2002 Salahun, Que´ffe´lec, and Tanne´
Downloaded 29 Mar 2002 to 193.54.246.80. Redistribution subject to AIP license or copyright, see http://ojps.aip.org/japo/japcr.jsp
4. is that relation ͑1͒ is only valid for thin plate samples in
which the dynamic demagnetizing fields are negligible in a
direction parallel to the planar surface of the material. The
demagnetizing factor along this direction is no longer equal
to zero for a 635-m-thick rectangular piece of LIFE com-
posite and depends of the ratio of the sample width on the
sample length.
Figure 5 shows that absorption is still important ͑Ͼ0.25͒
at high frequencies far from the magnetic losses region ͑1–4
GHz͒. This can be explained by the study of the field pattern
of the quasi-TEM mode. Microwave magnetic field compo-
nents perpendicular to the ferromagnetic layers exist in some
regions of the propagation structure. As these components
induce eddy currents, it results in a strong attenuation of the
propagating wave. The achievement of microwave stop-band
functions based on the gyromagnetic resonance requires a
lowering of the insertion losses out of the stop-band fre-
quency range. This is why we studied the influence of the
LIFE sample dimensions on the insertion losses in high fre-
quencies. The composite width must be optimized in order to
minimize magnetic field components orthogonal to the
planes of the CoNbZr layers. The ratio of the strip width W
to the sample width L is called d. When the central conductor
covers the whole sample width ͑dϾ 1͒, lower insertion losses
are obtained in high frequencies ͑Fig. 7͒. At 10 GHz, the
transmission response level is found to be lower than 3 dB in
spite of the impedance mismatch of the line due to the dis-
continuity that occurs between the foam substrate and the
LIFE material.
When the LIFE sample is larger than the central conduc-
tor ͑dϽ 1͒, the propagation wave is more attenuated. Indeed,
when dϭ0.75, S21 magnitude is on the order of 6 dB at 10
GHz ͑Fig. 7͒. The high value of d ensures that the micro-
wave electrical field is perpendicular to the ferromagnetic
layers over their whole width and microwave magnetic field
parallel to the layers. When Wϭ3.3 mm, the characteristic
impedance of the microstrip line without the LIFE sample
͑access lines͒ is 50 ⍀. Thus, L must be less than 3.3 mm to
verify the condition dϾ1. However, the chosen value for L
must not be too small to keep a high level of magnetic losses
in the stop band. From Fig. 7, it is obvious that the frequency
at which maximal absorption occurs depends on the sample
width. Indeed, the gyromagnetic resonance frequency is a
function of the demagnetizing factors which are related to
the ratio of the material width to the conducting strip width
as shown in the next section.
A typical band-stop function can be observed in Fig. 7
with 3 dB bandwidth of 4 GHz and maximum attenuation of
32 dB at a resonance frequency of 3.24 GHz. These experi-
mental results on the tunability of the gyromagnetic reso-
nance with dc field in the stripline and on the absorption of
the microwave signal in the microstrip line appear to be
promising for the achievement of magnetically tunable cir-
cuits.
III. THEORY
A. Electromagnetic analysis
In order to design a useful tunable device using the pre-
viously described propagation structure, an electromagnetic
analysis of the loaded microstrip line was performed. The
theory used is based on the work of Hines and Tsutsumi:16,17
it assumes that the center conductor of the line is substan-
tially wide compared with the thickness of the LIFE compos-
ite. This is the case for our propagation structure since the
ratio of the strip width to the substrate thickness is greater
than 5. Then, most of the microwave energy is confined be-
tween the strip and the lower ground plane. General solu-
tions to Maxwell’s equations are written for this center re-
gion which propagates a well-defined transverse electric
mode. This mode has well-known modal properties, with
field components Ey , Hx , and Hz . Ey is assumed to be
invariant with y. The differences between Tsutsumi’s struc-
ture and ours lie in the dc magnetic field direction and in the
geometry of the different layers of the microstrip substrate
͑Fig. 6͒. The propagation constants and the field patterns of
the various modes are determined by applying continuity
conditions for the tangential fields Ey and Hz at the bound-
aries xϭϮL/2 between the foam substrate and the LIFE ma-
terial, and at magnetic wall boundary Hzϭ0 at both edges of
the strip at xϭϮW/2.
Helmholtz’s equation for Ey is in the foam substrate
ץ2
Ey
d
ץx2
ϩ
ץ2
Ey
d
ץz2
ϩ2
00rEy
d
ϭ0, ͑5͒
FIG. 6. Optimized position of the LIFE sample in the microstrip line.
FIG. 7. Magnitude of the measured transmission coefficient as function of
frequency for various sample width.
5452 J. Appl. Phys., Vol. 91, No. 8, 15 April 2002 Salahun, Que´ffe´lec, and Tanne´
Downloaded 29 Mar 2002 to 193.54.246.80. Redistribution subject to AIP license or copyright, see http://ojps.aip.org/japo/japcr.jsp
5. and in the LIFE composite material:
ץ2
Ey
m
ץx2
ϩ
ץ2
Ey
m
ץz2
ϩ2
00effeffEy
m
ϭ0, ͑6͒
where 0 and 0 are, respectively, the permittivity and per-
meability of free space, r is the relative permittivity of the
foam substrate, and the radian frequency.
The solutions of Eqs. ͑4͒ and ͑5͒ are expressed as
Ey
d
ϭ͓Adsh͑kdx͒ϩBdch͑kdx͔͒exp͓i͑tϪ␥z͔͒ ͑7͒
Ey
m
ϭ͓Amsh͑kmx͒ϩBmch͑kmx͔͒exp͓i͑tϪ␥z͔͒, ͑8͒
with
kd
2
ϭ␥2
Ϫ2
00r ,
and
km
2
ϭ␥2
Ϫ2
00effeff ,
where Ad , Am , Bd , and Bm are arbitrary constants and ␥
represents the propagation constant in the z direction. Hz is
calculated from Ey using the Maxwell equation:
rot EϭϪj0effH. ͑9͒
The characteristic equation of the loaded microstrip line
is obtained by matching the electromagnetic fields Ey and Hz
at xϭϮL/2, and applying magnetic wall boundary Hzϭ0 at
xϭϮW/2. The dispersion relation is obtained by making the
determinant of the matrix form of the characteristic equation
to zero as
chͩkm .L
2 ͪ.shͩkm .L
2 ͪ.chͩkd .W
2 ͪ.shͩkd .W
2 ͪϭ0. ͑10͒
In order to validate the electromagnetic analysis of the struc-
ture, the calculated and measured transmission responses of a
microstrip line loaded with a 25.3ϫ3ϫ0.635 mm3
rectangu-
lar piece of LIFE material are compared in Fig. 8. For this
comparison only, the dominant mode of the line was taken
into account. Theoretical and experimental transmission re-
sponses of the line reveal a peak absorption of the micro-
wave energy due to the gyromagnetic resonance of the LIFE
material. It can be noticed that the agreement between theo-
retical and experimental data is quite satisfactory in a very
large frequency range. However, the theoretical curve in-
creases faster than the experimental one after the gyromag-
netic resonance frequency. The reason is that the Bloch–
Bloembergen’s theory used to compute the effective
permeability of the LIFE material was developed for single-
crystal ferrites. The amorphous nature of the CoNbZr layers
was not taken into account in our calculations.
B. Influence of the demagnetizing fields
The shift of the gyromagnetic resonance frequency as
function of the parameter d observed in Fig. 7 can be ex-
plained by dynamic demagnetizing effects. Indeed, dynamic
demagnetizing fields find their origin in the magnetic poles
generated by the microwave flux entering and exiting the
ferromagnetic layers.18
As no analytical relation describes
this phenomenon, electromagnetic simulations were realized
using the OPERA2D software19
to confirm this hypothesis.
Material length is neglected in this calculation because of the
quasi-TEM mode assumption used in the software. Calcula-
tions of the demagnetizing coefficients are based on the con-
servation of the magnetic flux. As the material width L de-
creases ͑d increases͒, the Ny coefficient decreases.
Demagnetizing fields along the Oy direction are less impor-
tant when d increases. Theoretical gyromagnetic resonance
frequency for a multilayered material is given by:20
frϭ␥ͱ͓H0ϩHaϩq.͑NxϪNz͒4MS͔.͓H0ϩHaϩq.͑NyϪNz͒4MS͔, ͑11͒
where Ny, Nx , and Nz are the demagnetizing factors in di-
rections perpendicular and parallel to the planar surface of
the plate material, respectively, and q is the volumic fraction
in ferromagnetic material. Figure 9 shows the comparison
between theoretical and experimental values for the gyro-
magnetic resonance frequency. The results are in close agree-
ment. The observed differences could be due to the static
demagnetizing field along the propagation direction or to the
nonflatness of ferromagnetic layers.
IV. EXPERIMENTAL RESULTS
A. Test device
A 25.3ϫ3ϫ0.635 mm3
rectangular piece of LIFE com-
posite was inserted into the foam substrate of the microstrip
line ͑Fig. 6͒. The total thickness of the LIFE sample is 0.635
mm. This LIFE sample is embedded in a 0.635-mm-thick
foam substrate by making a small hole in the substrate. Thin
layers of polypropylene are deposited on each side of the line
FIG. 8. Comparison between theoretical and experimental transmission re-
sponse of the loaded microstrip line.
5453J. Appl. Phys., Vol. 91, No. 8, 15 April 2002 Salahun, Que´ffe´lec, and Tanne´
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6. substrate to hold the LIFE sample in its hole ͑Fig. 6͒. These
polypropylene layers facilitate the deposition of the conduc-
tive strip and of the line ground plane.
The S-parameter measurements of the line were per-
formed in the 130 MHz–10 GHz frequency band with a
network analyzer system ͑HP8720A͒. Systematic errors
caused by defects of the network analyzer ͑coaxial cables,
electronic components, etc.͒ were reduced by performing the
conventional short-open-load-through calibration. The im-
pedance mismatch at the coaxial-to-microstrip connection
was characterized and removed by exploiting the
S-parameter measurements of a 50-ohm foam-based micro-
strip line without the LIFE material ͑reference line: hϭ0.635
mm, Wϭ3.3 mm, rϭ1.07, and lengthϭ50 mm͒. Thus, the
studied transmission coefficient is in fact the forward S21
scattering term of the tunable circuits referenced to the char-
acteristic impedance of a LIFE-material-free microstrip line.
As a consequence, insertion losses are free from impedance
mismatch at the ends of the microstrip line.
The easy axis of CoNbZr films is parallel to the wave
propagation direction. In this configuration, the microwave
magnetic field hrf of the quasi-TEM mode is directed in the
transverse direction of the magnetic moments. This permits a
strong interaction between the wave and ferromagnetic lay-
ers. The static magnetic field H0 is applied parallel to the
strip of the line, i.e., along the easy axis of the CoNbZr films,
perpendicular to the microwave field hrf .
Two phenomena can be exploited to generate tunable
stop-band effects from the circuit previously described. The
absorption of the microwave signal occurring in the vicinity
of the gyromagnetic resonance frequency is exploited to
work out a tunable stop-band function, and then the in-plane
anisotropy of ferromagnetic thin films is used to develop a
magnetic switch.
B. Magnetic tunable stop-band function
To demonstrate the tunability of a microstrip line that
incorporates a LIFE sample, an external dc magnetic field is
applied along the wave propagation direction using an elec-
tromagnet. The magnitude of the measured transmission co-
efficient S21 as a function of frequency is shown in Fig. 10͑a͒
for a 50-mm-long microstrip line. The field strength applied
varies from 0 to 250 Oe. We can observe in Fig. 10͑a͒ that
the absorption peak in transmission generated by the gyro-
magnetic resonance phenomenon which occurs in the
CoNbZr films can be tuned from 3.1 GHz at zero field to 5.4
GHz at 250 Oe bias: This corresponds to a tunability of 54%.
In comparison, when a YIG ferrite sample is inserted in a
50-ohm-microstrip line, a tunability of about 32% is obtained
͓see Fig. 10͑b͔͒. This tunability is much lower than those
exhibited by the microstrip line containing the LIFE sample.
However, for spinel ferrites such as LiZn or NiZn with mag-
netization values about 5 kG, the difference between tun-
abilities is decreased. The stop-band depth exhibited by the
line that incorporates a LIFE sample is of about 30 dB at-
tenuation; this should be compared to less than 25 dB for the
line containing the YIG sample. The insertion loss of about 3
dB in high frequencies is primarily due to the discontinuities
that occur between the access lines and the central region
containing the LIFE sample. This could be improved by nar-
rowing the width of the center-strip conductor above the
sample as mentioned herein. The bandwidth of the absorp-
tion curves enlarges when the dc magnetic field increases.
The center frequencies defined as the frequencies where
maximal absorption occurs are shown in Fig. 11 ͑dots͒ as a
function of the applied field. The behavior observed is re-
lated to the gyromagnetic phenomenon, for which the reso-
nant frequency is given by Kittel’s formula.11
The theoretical
curve shown in Fig. 11 ͑linear curve͒ is in good agreement
with the center frequencies measured when values of 0.0028
and 0.83 for q(NxϪNz) and q(NyϪNz) are introduced in
Ref. 11. As previously mentioned, Nx , Ny , Nz are calculated
thanks to the OPERA2D software. To improve the perfor-
FIG. 9. Gyromagnetic resonance frequency as function of dϭW/L. Com-
parison between theoretical and experimental values of the resonance fre-
quency.
FIG. 10. Magnitude of the measured transmission coefficient as function of
frequency for various dc field strengths. ͑a͒ A LIFE sample is inserted in a
microstrip line. ͑b͒ A YIG ferrite is inserted in a microstrip line.
5454 J. Appl. Phys., Vol. 91, No. 8, 15 April 2002 Salahun, Que´ffe´lec, and Tanne´
Downloaded 29 Mar 2002 to 193.54.246.80. Redistribution subject to AIP license or copyright, see http://ojps.aip.org/japo/japcr.jsp
7. mances of the band-stop function, e.g., quality factor, the
ferromagnetic resonance linewidth ͑⌬H͒ of the ferromag-
netic thin films has to be decreased.
C. Magnetic switch
A magnetic switch can be worked out with the micros-
trip line previously described. Indeed, in Fig. 10, one should
notice that the magnitude of the transmission coefficient at
3.1 GHz rises from Ϫ32 to Ϫ4 dB when the dc magnetic
field ranges from zero to 250 Oe. Figure 12 depicts the trans-
mission response of a magnetic switch realized when the dc
magnetic field is applied in the x direction, i.e., perpendicular
to the easy axis of the ferromagnetic layers. At zero field, a
gyromagnetic absorption occurs because the magnetic mo-
ments are perpendicular to the microwave magnetic field hrf .
As a consequence, the transmission coefficient S21 is dis-
turbed by important insertion losses in the 2–4 GHz fre-
quency band. The attenuation level is of about 25 dB at 2.4
GHz. When a dc magnetic field with a sufficient strength is
applied transversally to the microstrip line, no gyromagnetic
phenomenon occurs because the magnetic moments are par-
allel to the microwave magnetic field. Then, the magnitude
of the transmission coefficient highly increases at low fre-
quencies. The strength of the dc field must be larger than the
sum of the ferromagnetic layers anisotropy field ͑Haϭ33 Oe͒
and of the static demagnetizing field. A dc magnetic field
strength of 200 Oe is required to obtain the transmission
response given in Fig. 12. In the 2–4 GHz frequency band,
the incident wave propagated along the microstrip is strongly
attenuated at zero field, whereas at 200 Oe, it propagates
with low absorption in the device ͉͑S21͉ϭϪ0.5 dB͒. The
resonance frequency can be further increased by superimpos-
ing a dc magnetic field along the easy axis.
V. CONCLUSION
A ferromagnetic composite was integrated in the sub-
strate of microstrip lines with respect of illumination condi-
tions to insure high intrinsic impedance of the material at
microwave frequencies. The feasibility of magnetically tun-
able stop-band devices using this composite material was
demonstrated. A tunable stop-band function based on the
shift of the gyromagnetic resonance frequency with a dc
magnetic field was realized. A magnetic switch using the
anisotropy of the ferromagnetic layers was performed. The
greatest interest of the use of a LIFE material in microstrip
circuitry is the high tunability which can be reached with dc
magnetic fields of low strength. Indeed, a 54% tunability was
obtained for the stop-band function with a 250 Oe field. For
a given dc field, the operating frequencies are much higher
than those obtained when a YIG ferrite is inserted in a mi-
crostrip line. This way of integrating a composite material in
guided wave structures opens the road to a generation of
low-cost tunable planar circuits based either on the gyromag-
netic resonance or on the dc field dependence of the perme-
ability. However, to be useful for serious microwave appli-
cations the electromagnetic properties of the composite, in
particular the resonance linewidth, have to be improved.
ACKNOWLEDGMENT
The authors are grateful to S. Pinel, B. Della, and Dr. C.
Person for their contribution toward the realization of the
microstrip structures. They also thank Dr. J. L. Matte´i for his
help during the simulations with the OPERA2D software. All
of them are members of the LEST.
1
G. F. Dionne and D. E. Oates, IEEE Trans. Microwave Theory Tech. 44,
1361 ͑1996͒.
2
H. How, T. M. Fang, and C. Vittoria, IEEE Trans. Microwave Theory
Tech. 43, 1620 ͑1995͒.
3
D. E. Oates and G. F. Dionne, IEEE MTT-S Digest 1, 303 ͑1996͒.
4
M. Tsutsumi and T. Fukusako, IEEE MTT-S Digest 3, 1491 ͑1997͒.
5
I. Huynen et al., IEEE Trans. Microwave Guided Wave Lett 9, 401
͑1999͒.
6
C. S. Tsai, J. Su, and C. C. Lee, IEEE Trans. Magn. 35, 3178 ͑1999͒.
7
O. Acher et al., IEEE Trans. Microwave Theory Tech. 44, 674 ͑1996͒.
8
G. Perrin et al., Le vide: science, technique et applications 275, 256
͑1995͒.
9
M. Born and E. Wolf, Principle of Optics ͑Pergamon, New York, 1959͒, p.
803.
10
N. Bloembergen, Phys. Rev. 78, 572 ͑1950͒.
11
E. Salahun et al., IEEE Trans. Magn. 37, 2743 ͑2001͒.
12
G. Bartollucci and R. Marcelli, J. Appl. Phys. 87, 6905 ͑2000͒.
13
M. Pardavi-Horvath, J. Magn. Magn. Mater. 215, 171 ͑2000͒.
14
H. Lattard, C. Person, J. P. Coupez, and S. Toutain, France Patent No.
13427 ͑1997͒.
15
HP-MDS Software, release 7.10.
16
M. E. Hines, IEEE Trans. Microwave Theory Tech. 19, 442 ͑1971͒.
17
M. Tsutsumi and K. Okubo, IEEE Trans. Magn. 28, 3297 ͑1992͒.
18
G. F. Dionne and D. E. Oates, J. Appl. Phys. 85, 4856 ͑1999͒.
19
OPERA 2D, Vector Field.
20
L. Adenot, Ph.D. dissertation, University of Bordeaux, 2000.
FIG. 11. Comparison of the measured and calculated stop-band center fre-
quency as function of the applied static magnetic field.
FIG. 12. Transmission response of the magnetic switch.
5455J. Appl. Phys., Vol. 91, No. 8, 15 April 2002 Salahun, Que´ffe´lec, and Tanne´
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