EMC ROME 1996: EVALUATION OF RADIATED EMISSIONS FROM PCB AND CABLES AT POST-LAYOUT LEVEL

1. EVALUATION OF RADIATED EMISSIONS FROM PRINTED
CIRCUIT BOARDS AND CABLES AT POST-LAYOUT LEVEL
E. Leroux
High Design Technology
Corso Trapani, 16
10139 Torino Italy
S. Caniggia
ITALTEL
Castelletto di Settimo
Milanese (MI) 20019 Italy
F. Canavero
Politecnico di Torino
Corso Duca degli Abruzzi,
24-10100 Torino Italy
B. Demoulin
University of Lille
59655 Villeneuve d’ascq
France
Abstract - Because it costs to solve
ElectroMagnetic Compatibility (EMC) problems late
in the development process, new methods have to
predict radiated electromagnetic emissions at the
design stage. In the case of complex Printed Circuit
Boards (PCBs) with attached cables, a tradeoff
between accuracy and simulation time must be found
for this evaluation. In this paper the Radiated
Emissions (RE) of single boards with and without an
attached cable are investigated. A feasibility study is
proposed for the modelling of these emissions. The
results are compared with some measurements for
validation. Some ideas are given to improve the
models and to introduce them into an EMC post-
layout environment.
1. INTRODUCTION
Since January 1996, electronic products have also to
pass the requirements of the european norms in
ElectroMagnetic Compaltibility (EMC) to obtain the
EEC mark and be sold in the european market. In
many cases electronic equipment is mostly composed
of Printed Circuit Boards (PCBs), so PCB's
manufacturers will have properly to limit the
Radiated Emissions (RE) of the boards they produce.
Currently, the most common method of handling
ElectroMagnetic (EM) emissions is through
compliance testing of the first prototype, already
implemented.
In the case of a PCB it would be necessary to have
the first board made by the manufacturer, and if it
does not pass the tests, repeat again all the
manufacturing processes. And this may delay the
product's completion date and increase the unit cost
of the product because the designer has not as many
options available for correcting an EMC problem
late in the development process. A solution to limit
this possible iteration is to predict the EM Field at
the design stage, thanks to a computer-oriented
analysis of EM radiated Interferences (EMI).
This proposal seems to be quite hard to solve
because of the number of factors that influence the
radiated EM field. As the microstrip structures that
compose the nets play a major role especially in the
frequency bands of the norms, a method [1] has been
developped to predict their differential mode
emissions. This algorithm uses the dyadic Green's
functions of the actual medium (the PCB) and is able
to take into account of the dielectric layers in the
field calculation. In particular dielectric covers
modify the radiation spectrum of PCBs and have to
be considered in a predictive method. But the
presence of unintended common mode currents on
the external cables of electronic equipment is often
the primary source of radiated ElectroMagnetic
Interferences (EMI) at frequencies above 30 MHz. In
this paper a feasibility study on the modelling of RE
of simple two-layered PCBs with and without an
attached cable is presented. Simulations and
measurements are conducted on two simple test
boards made of a microstrip and a stripline. Some
ideas are given to improve the models and to
introduce them into an EMC post-layout
environment.
2. DESCRIPTION OF THE COUPLING AND
THE TEST BOARDS
A metal plane is commonly thought of as a low
impedance path for returning currents and one
which is of constant potential across its area. In fact,
these currents give rise to voltage gradients in the
plane which act as sources of common-mode
current. The majority of the RF current flowing
along a signal trace on a multi-layer PCB returns on
the ground plane directly beneath the signal trace.
However, a small portion of the ground-plane
current also can return via indirect paths causing the
PCB and attached cables to produce common-mode
radiation similar to a dipole antenna. To investigate
the effect of these radiation sources in actual PCBs,
two structures are considered (see Figure 1): one
wire or strip over a ground plane separated by
dielectric (microstrip structure), one wire or strip
between two ground planes with dielectric (stripline
structure). Both the structures were tested with and
without an attached cable.
Models were built to simulate the radiation of these
two PCBs with attached cables powered by the PCB
ground noise.
In order to verify the models, some experiments
were conducted. For every structure a 8-MHz
oscillator, with output resistance of 50 W, was
mounted on a board enclosed in a shielded box of

2. size: 5.5´5´2.5 cm (see Figure 2). The output
connector was a SMA type. This provided a 8 MHz
trapezoidal pulse train having roughly a 50-percent
duty cycle, a rise and fall time of approximately 2 ns
and amplitude of 5 V without load. A battery-
powered supply consisting of a 9 V battery and a
7805 regulator provided the 5 V DC for the
oscillator. It is important to note that we never used
connection to the commercial power system.
M icro strip
S trip line
h
2 h
P lanes: W id th= 1 9 cm L ength= 3 0 cm
er= 4 .4 w ire: rad ius= 0 .5 m mh= 1 .6 m m
Z 0 = 1 5 3 W
Z 0 = 1 2 5 W
Figure 1: Examined structures for multilayer circuit
boards: microstrip and stripline.
Shielded box
PCB local
plane
tfall=2 nsec
Period of 125 nsec
trise=2 nsec 5 Volts
Source: Digital gate AC 244
50W
PCB track 100 W
8Mhz
Figure 2: The 8 MHz digital circuit with the output
waveform used in shielded box
The models use the envelope of the generator output
spectrum. In Figure 3, a comparison between the
measured spectrum of the source on the 50W input
of a spectrum analyser and the envelope of the
calculated spectrum is given.
1 10
7
1 10
8
1 10
950
60
70
80
90
100
110
120
Figure 3: Generator output spectrum on a 50 W load.
For the calculations reported in this paper, we have
always used the spectral bounds minus 3 dB to take
into account the rms values measured by the
spectrum analyser. In this paper the radiated
emissions were measured at 3 meters of distance
with the antennas at 1.1 m height from the metallic
ground floor, in horizontal polarisation. In order to
maximise the emission the PCB was orientated with
the plane in vertical position and the center of the
PCB was 1.1 m above the ground plane of the
chamber. The PCB track was terminated with a 100
W resistance.
3. THE MICROSTRIP STRUCTURE
3.1 Differential mode emissions of the PCB alone
Firstly the differential mode emissions of the PCB
are investigated by means of measurements and
simulations.
For the microstrip structure, as no dielectric cover is
present, the Hertzian Radiating Dipoles Method [2]
has been used to model the differential mode
emissions:
· the current distribution on the track is obtained
using the Transmission Line Theory (TLT) from
the envelope of the generator calculated
spectrum
· the track is divided into segments for which the
lenght L << l, l is the considered wavelenght
· the classical elementary Hertzian dipole formula
is used to calculate the emissions of each
segment
· the PCB local plane and the floor of the semi-
anechoic chamber are taken into account using
image theory
The results are reported in Figure 4. Note that the
calculated differential mode emissions fit quite well
with the higher measured results.
Frequency in MHz
|V| in dBmV
envelope of the
calculated spectrum
measured spectrum

3. 10 100 1 10
320
10
0
10
20
30
40
50
60
70
80
EN55022 norm
simulation
measurement
frequency in MHz
|E| in dBmV/m
Figure 4: Microstrip configuration: RE of only PCB
3.2 Modelling of common mode currents on the
PCB
Secondly, a more accurate model to predict radiated
emission from a microstrip structure is presented
(Figure 5). In this figure 5, one can observe that the
total radiated field is the sum of two contributions: a
differential mode radiation EDM due to the
differential signal current Isign; a common mode
radiation ECM due to the ground noise Vn always
present in a PCB with a finite dimensions ground
plane.
IDM» +
Etot=EDM+ECM ECM
EDM
ICM
Vn
Ltrans= Lplane - Mplane/track
Isign»IDM
Ground
noise Vn
Vn=j w Ltrans Isign
Etot,EDM,ECM are vectorial quantities
Figure 5: Radiated emission from a PCB as sum of
common mode (ground noise) and differential mode
(signal current).
This ground noise depends on the coupling between
the track, the PCB local plane and the metal floor of
the semi-anechoic chamber. The finite ground plane
of a PCB has an associated partial inductance Lgnd
[3]. But it is not only the self-inductance of the
plane that matters but also its coupling to the (much
larger) inductance of the track [4].
Together these parameters determine the voltage
drop over the PCB local plane, if a current flows
trough the track and returns through the plane. This
voltage drop is proportional to the signal differential
current Isign and is responsible for common mode
emission through the distributed stray capacitance of
the ground plane or by a cable attached to the PCB
(see [4] [5]). In absence of an I/O cable, the
contribution of the common mode radiation ECM
could also be neglected in the case of a PCB having
a large ground plane and placed in vertical position.
This is not the case when a cable is attached to the
PCB.
3.3 Emissions of the PCB with an attached cable
3.3.1 The used model
Finally the emissions of the PCB with an attached
cable are investigated. A cable of 1 m is attached to
the structure of Figure 1, in horizontal position and
at 1.1m from the ground of the chamber floor, it is
not connected at the other extremity (high
impedance). The model of Figure 6 has been used. It
has already been explained in [4] and especially in
[6] in which the feasibility study started. In the
present article some improvement respect to [6] are
presented.
cable
Vn=j wLtrans Isign
C10
metal floor of semi-anechoic room
Ltrans= Lplane -Mplane/track
H
Figure 6: The used model for the cable exitation
The modelling follows three steps:
· The determination of the ground noise.
C10 is the capacitance between the PCB local plane
and the metal floor of the semi-anechoic chamber.
Ltrans is a type of a “transfer inductance”, it depends
on the mutual inductance between the PCB local
plane and the track. Respect to [6] these elements
have not been calculated using 2D approximations
or analytical formula but using the Partial Element
Equivalent Circuit (PEEC) method (see 3.3.2). A
purpose of this article is to see if the prediction of
the radiated emissions can be by this way improved
· The propagation of the ground noise on the
cable.

4. The Transmission line theory (TLT) is used to
calculate the current distribution at any abscissa on
the cable. The distance H between the PCB local
plane and the floor of the semi-anechoic chamber
defines the maximum frequency for which TLT can
approximatively apply:
fmax
.
=
3108
l
with l = H =1.3 meters, fmax = 230
Mhz
This frequency correspond to the maximum
frequency for which the emissions of the cable is
much bigger than the ones of the PCB (see 3.3.3). In
fact the method is used until this maximum
frequency.
· The calculation of the cable emissions.
The Hertzian Radiating Dipoles Method has been
used dividing the cable in short segments to
calculate the RE of the cable. The floor of the semi-
anechoic chamber is taken into account using image
theory
3.3.2 The use of PEEC method to determine the
concentrated desired elements
The PEEC method allows the evaluation of parasitic
parameters in threedimensional structures of
conductors and was introduced in 1974 by Ruehli
[7] [8]. The key idea is to subdivide all conductors
in volume cells where conduction or polarization
current flows and in surface cells where free or
boundary charge is located. Inductance and
coefficient of potential matrices are then calculated,
whose elements are called partial, as they are
referred to the single cell and not to the whole
conductor. Calculated values can be directly used as
circuit parameters (inductances and capacitances),
or can be grouped in order to obtain macromodels,
as it has been done in this application.
This method allows to model very general
structures, because the only restriction in choosing
cells is due to their shape: their size must be parallel
to cartesian axes. Using PEEC approach it is
possible to study PCBs placed either parallel or
perpendicular to the floor of the semianechoic
chamber, only changing cells layout.
In this paper, the interaction between the microstrip
(microsystem) and the floor of the semianechoic
chamber (macrosystem) has been evaluated.
In Figure 7 the subdivision of the structure in
volume conductive cells to evaluate the inductive
coupling is represented.
Figure 7: Example of discretization in volume cells
(3 for the PCB local plane and 1 for the PCB track).
The quantity Ltrans=Lplane - Mplane/track, was to be
evaluated: it has not been necessary to introduce
cells on the floor. Furthermore the current mainly
flows parallel to the track, then volume cells were
placed only along this direction, and not
transversally.
Repeated tests leaded to the convergency of Ltrans at
a value of 0.9 nH, using 25 volume cells in both
conductors. With the analytical formula in [6], Ltrans
was equal to 0.89 nH.
The evaluation of the capacitive coupling has been
done without modeling dielectric: C10 (coupling
between the PCB local plane and the floor) was the
only quantity of interest, representing the coupling
between the micro and the macrosystem.
Repeated tests leaded to the convergency of C10 at a
value of 5.27 pF, using a grid of 5x5 uniform cells
on the track and 7x7 on the local plane and on the
floor. A value of 10 pF has been obtained with the
2D approximation in [6].
3.3.3 Comparison between simulation and
measurement
The Figure 8 shows the results of simulation of the
emissions from the PCB alone, the attached cable,
the PCB with the attached cable and the
measurement of the emissions from the PCB with
the attached cable.

5. 10 100 1 10
320
10
0
10
20
30
40
50
60
70
80
FrequencyinMHz
|E| indBmV/m
EN55022
EmissionsfromPCB+
cable: measurement
EmissionsfromPCB+
cable: simulation
EmissionsfromPCB:
simulation
Emissions fromthe
cable: simulation
8
Figure 8: Microstrip configuration. Simulation of
the emissions from the PCB alone, the attached
cable, the PCB with the attached cable.
Measurement of the emissions from the PCB with
the attached cable
The comparison of the experimental results of
Figure 4 and 8 shows that the cable has a major
contribution on the RE from the system PCB +
cable mainly below 250 Mhz. In deed, the cable is
one meter long and the PCB is only 30 cm long.
Until fmax = 230 MHz the maximum frequency for
which it is possible to apply the method, the
simulated envelope follows reasonably the measured
spectrum. But the most significant frequency is
shifted and the method underestimates the maximum
radiated field.
It is probably due to the use of the envelope of the
signal injected on the track and the fact that the
method does not take into account of the coupling
by proximity between the track and the cable.
The use of a 3D method to calculate the parasitic
elements C10 and Ltrans does not improve drastically
the results of the radiated emissions in dB respect to
[6] as the inductive effect is predominant and was
predicted in a good approximation in [6] (see 3.3.2).
The quantity Ltrans does not depend on the presence
of the ground floor of the semi-anechoic chamber, it
can be so approximatively determined by the use of
analytical formula also for a PCB placed in a
vertical position. For non terminated lines the
capacitive effects have a greater influence. As the
PEEC method gives a 3D evaluation of C10, it is
worth using it to determine the capacitive coupling
when the board is placed in horizontal or vertical
position.
4. THE STRIPLINE STRUCTURE
4.1 Emissions of the PCB alone
Respect to [6] an other very common PCB structure
has been examined in this paper: it is the stripline
type that can be found in multilayer PCBs with
several ground planes. Due to the nature of the
structure, one wire/strip between two ground planes,
one could expect that the ground planes act like a
shield and no practical emission should be
measured. This is true if the planes are tiedly
connected together or, in the limit, also attached
around their periphery in order to form a perfectly
closed container. But, if one plane is floating or
badly connected to the other plane, resonances
occur and produce emissions higher than those
produced by an equivalent microstrip structure [9]
In our case of study the two planes are tiedly
connected. One must so realise that the structure is
quasi-symmetric, the signal current is quasi-equally
divided between the two planes and the emission is
only due to a small percentage difference between
the current of the two ground planes. The current
distribution on the ground planes depends on the
frequency. But for striplines used in modern high
speed digital electronics, the currents on the
structure can be calculated by a multiconductor
transmission line model taking into account that the
return current in the ground plane is not spread more
than 4-5 times the distance between the strip and
one of the ground plane [10]. The equivalent model
consists of three conductors: one for the track and
the other two for the planes spaced a distance 2 h as
shown in Figure 1. If a signal differential current
Isign spreads along the conductor which models the
track, a current Isign (1+K)/2 spreads in the
conductor which models one metal plane, a current
Isign (1-K)/2 spreads in the conductor which models
the other plane.
For example, the calculated values reported in
Figure 9 were obtained by a difference of about
20% between the two return currents on the ground
planes (K=0.2) and take into account the phase
differences.

6. 10 100 1 10
320
10
0
10
20
30
40
50
60
70
80
|E| in dBmV/m
EN 55022
simulation
measurement
Frequency in Mhz
Figure 9: Stripline configuration: emissions of the
PCB alone
The simulated envelop follows reasonably the
measured spectrum. But it would be necessary to
justify better the difference of about 20% between
the two return currents on the ground planes.
4.2 Emissions of the PCB with an attached cable
4.2.1 The used model
A cable is attached to one of the two planes of the
stripline. The same model as in 3.3.1 is used but the
equivalent generator Vn is equal to: j w Ltrans Isign /2.
The quantities Ltrans and C10 have to be calculated
for a stripline structure placed in vertical position
above the metal floor of the semi-anechoic chamber.
4.2.2 The use of PEEC method to determine the
concentrated desired elements
The same approach as in 3.3.2 is used to determine
the capacitive coupling between a stripline in
vertical position and the metal floor of the chamber.
Repeated tests leaded to the convergency of C10 at a
value of 10.7 pF, using a grid of 3x3 uniform cells
on the track and 9x9 on the local plane and on the
floor.
As far as inductive coupling is concerned, Ltrans has
been evaluated by a closed form formula.
4.2.3 Comparison between simulation and
measurement
The Figure 10 shows the results of simulation of the
emissions from the PCB alone, the attached cable,
the PCB with the attached cable and the
measurement of the emissions from the PCB with
the attached cable.
10 100 1 10
320
10
0
10
20
30
40
50
60
70
80
|E| in dBmV/m
Frequency in MHz
EN 55022
emissions fromthe PCB +
cable: simulation
emissions from the PCB
+ cable: measurement
emissions fromthe
PCB alone
emissions from the
cable: simulation
Figure 10: Stripline configuration. Simulation of the
emissions from the PCB alone, the attached cable,
the PCB with the attached cable. Measurement of
the emissions from the PCB with the attached cable.
With a cable attached to one of the two planes of the
stripline structure the same level of emission as
those obtained with the microstrip structure is
noticed. The same remarks presented for the
modelling of the microstrip structure apply for the
stripline.
CONCLUSION
A feasibility study on the modelling of the Radiated
Emissions (RE) from simple two-layered PCBs with
and without an attached cable has been presented.
The validity domain of these models has been
evaluated by the means of comparisons with
measurements.
A numerical approach as the PEEC method is
necessary to determine the capacitive coupling
between the PCB and the metal floor of the semi-
anechoic chamber especially when the board is in
vertical position. The inductive coupling can be
rapidly determined by the use of close-form
formulas but it is more accurately determined using
PEEC approach.
A major improvement on the presented modelling
could be the use of Simulation Program of Response
of Integrated Network Transients (SPRINT) [11]
simulator to obtain the current on the track instead
of using the envelope of the calculated spectrum at
the near end of the PCB track. As SPRINT is
embedded in an EMC post-layout environment:
Post-layout Rapid Exhaustive Simulation and Test of
Operation (PRESTO) [11], these models could be
inserted in PRESTO in order to complete the
PRESTO/EMIR solution for RE prediction.

7. In the presented examples the cable was responsible
for non passing the EN 55022 limits. The cable is a
radiating structure which is powered by the PCB
ground noise. Reducing the ground noise at the PCB
level will reduce the emissions level of the cable. It
is so useful to use an EMC environnment for PCB to
try to control the PCB ground noise already at Post-
layout level.
A stripline, of which the two planes are tiedly
connected, is a quite symmetrical structure and
produces low emissions. But with a cable attached
to one of the two ground planes the stripline radiates
as a microstrip with an attached cable.
ACKNOWLEDGEMENTS
The authors thank C. Giachino, A. Giuliano for their
help to use the PEEC approach and L. Vitucci, G.
Marano for their participation to the measurements
in ITALTEL.
REFERENCES
[1] E. Leroux, F. Canavero, G. Vecchi, "Prediction
of radiated electromagnetic emissions from PCB
traces based on Green dyadics," Proc. EURO-DAC,
Brighton (UK), Sept. 18-22, 1995, pp. 354-359.
[2] S. Caniggia, L. Catello, V. Costa, “On the
computation of electromagnetic filed components
fron a transmission line: theory and application”,
IEEE 1990 International Symposium on
Electromagnetic Compatibility, Washington USA,
21 - 23 August 1990
[3] T. H. Hubing, T .P. Van Doren, J. L. Drewniak,
“Identifying and Quantifying Printed Circuit Board
Inductance”, IEEE International Symposium 1994
[4] J. R. Bergervoet, “EMC Measurements and
Models Connecting the System Level with the
Module Level”, Philips Journal of Research, Vol.
48, pp.63-81, 1994
[5] R. W. Dockey, R. F. German, “New Techniques
For Reducing Printed Circuit Board Common-Mode
Radiation”, IEEE International Symposium 1993
[6] E. Leroux, F. Canavero, S. Caniggia, B.
Demoulin, “Contribution des méthodes analytiques
pour la prédiction du rayonnement de cartes et
connectique”, International Symposium and
exhibition on EMC, 2 - 5 september 1996, Lille
France
[7] A. E. Ruehli, “Equivalent circuit models for
three-dimensional multiconductor systems”, IEEE
Trans. on Microwave Theory and Techniques, v.
MTT-22, pp. 216-221, march 1974
[8] A. E. Ruehli, P. A. Brennan, “Efficient
capacitance calculations for three-dimensional
multiconductor systems”, IEEE Trans. on
Microwave Theory and Techniques, v. MTT-21, pp.
76-82, february 1973
[9] S. Caniggia, L. Catello, V. Costa, G. Marano, N.
O’Riordan, L. Vitucci, “Controlling radiated
emission from a physically large telecommunication
system”, paper to be published on IEEE Trans. on
EMC, August 1996
[10] S. Celozzi, A. Orlandi, “PCB emission analysis
in different image planes configurations”,
International Wroclaw 1994 symposium on EMC,
Wroclaw Poland
[11] S. Forno, S. Rochel, "Advanced simulation and
modeling techniques for hardware quality
verification of digital systems," Proc. EURO-DAC,
Grenoble (F), 1994.