1. A Phase Shifted Semi-Bridgeless Boost Power Factor
Corrected Converter for Plug in Hybrid Electric
Vehicle Battery Chargers
Fariborz Musavi
Department of Research, Engineering
Delta-q Technologies Corp.
Burnaby, BC, Canada
fmusavi@delta-q.com
Abstract—In this paper, a phase shifted semi-bridgeless boost
power factor corrected converter is proposed for plug in
hybrid electric vehicle battery chargers. The converter features
high efficiency at light loads and low lines, which is critical to
minimize the charger size, charging time and the amount and
cost of electricity drawn from the utility; the component count,
which reduces the charger cost; and reduced EMI. The
converter is ideally suited for automotive level I residential
charging applications.
A detailed converter description and steady state operation
analysis of this converter is presented. Experimental results of
a prototype boost converter, converting universal AC input
voltage to 400 V DC at 3.4 kW are given and the results are
compared to an interleaved boost converter to verify the proof
of concept, and analytical work reported. The results show a
power factor greater than 0.99 from 750 W to 3.4 kW, THD
less than 5% from half load to full load and a peak efficiency of
98.6 % at 240 V input and 1000 W load.
I.
INTRODUCTION
A plug in hybrid electric vehicle (PHEV) is a hybrid
vehicle with a storage system that can be recharged by
connecting a plug to an external electric power source. The
charging AC outlet inevitably needs an on-board AC-DC
charger with power factor correction [1]. An on-board 3.4
kW charger could charge a depleted battery pack in PHEVs
to 95% charge in about four hours from a 240 V supply [2].
A variety of circuit topologies and control methods have
been developed for PHEV battery chargers. The two-stage
approach with cascaded PFC AC-DC and DC-DC converters
is the common architecture of choice for PHEV battery
chargers, where the power rating is relatively high, and
lithium-ion batteries are used as the main energy storage
system [3]. The single-stage approach is generally only
suitable for lead acid batteries due to large low frequency
ripple in the output current.
This work has been sponsored and supported by Delta-q Technologies
Corporation.
1
Wilson Eberle and 2 William G. Dunford
Dept. of Electrical and Computer Engineering
University of British Columbia | 1 Okanagan | 2 Vancouver
1
Kelowna, BC, Canada | 2 Vancouver, BC, Canada
1
wilson.eberle@ubc.ca | 2 wgd@ece.ubc.ca
In the two-stage architecture, the PFC stage rectifies the
input AC voltage and transfers it into a regulated
intermediate DC link bus. At the same time, power factor
correction is achieved [4]. The boost circuit-based PFC
topology operated in CCM is employed in this study as the
main candidate for front end single-phase solutions for ACDC power factor corrected converters used in PHEV battery
chargers.
A. Conventional Boost Converter
The conventional boost topology shown in Fig.1 uses a
dedicated diode bridge to rectify the AC input voltage to DC,
which is then followed by the boost section. In this topology,
the output capacitor ripple current is very high [5] and is the
difference between diode current and the dc output current.
Furthermore, as the power level increases, the diode bridge
losses significantly degrade the efficiency, so dealing with
the heat dissipation in a limited area becomes problematic.
Due to these constraints, this topology is good for a low to
medium power range up to approximately 1kW. For power
levels >1kW, typically, designers parallel semiconductors in
order to deliver greater output power. The inductor volume
also becomes a problematic design issue at high power
because of permeability drops at higher load and heat
associated with core and copper losses.
Figure 1. Conventional PFC boost topology

2. B. Bridgeless Boost Converter
The bridgeless configuration topology shown in Fig.2
avoids the need for the rectifier input bridge yet maintains
the classic boost topology [6-13]. It is an attractive solution
for applications >1kW, where power density and efficiency
are important. The bridgeless boost converter solves the
problem of heat management in the input rectifier diode
bridge, but it introduces increased EMI [14, 15]. Another
disadvantage of this topology is the floating input line with
respect to the PFC stage ground, which makes it impossible
to sense the input voltage without a low frequency
transformer or an optical coupler. Also in order to sense the
input current, complex circuitry is needed to sense the
current in the MOSFET and diode paths separately, since the
current path does not share the same ground during each
half-line cycle [8, 16].
Figure 2. Bridgeless PFC boost topology
C. Interleaved Boost Cconverter
The interleaved boost converter shown in Fig.3 is simply
two boost converters in parallel operating 180° out of phase
[20-22]. The input current is the sum of the two inductor
currents. Because the inductors ripple currents are out of
phase, they tend to cancel each other and reduce the input
ripple current caused by the boost switching action. The
interleaved boost converter has the advantage of paralleled
semiconductors. Furthermore, by switching 180° out of
phase, it doubles the effective switching frequency and
introduces smaller input current ripple, so the input EMI
filter can be smaller [23-25]. This converter also has reduced
output capacitor high frequency ripple, but it still has the
problem of heat management for the input diode bridge
rectifiers.
Figure 3. Interleaved PFC boost topology
In the following section, a new phase shifted semibridgeless boost PFC converter is proposed in order to
improve overall efficiency of the AC-DC PFC converter,
while maintaining all the advantages of the existing
solutions.
II.
PHASE SHIFTED SEMI-BRIDGELESS BOOST TOPOLOGY
The phase shifted semi-bridgeless topology shown in
Fig.4 is proposed as a solution to address the problems
outlined in section I for the conventional boost, bridgeless
boost and interleaved boost topologies. This topology
features high efficiency at light loads and low lines, which is
critical to minimize the charger size, charging time and the
amount and cost of electricity drawn from the utility; the
component count, which reduces the charger cost; and
reduced EMI. The converter is ideally suited for automotive
level I residential charging applications in North America
where the typical supply is limited to 120V and 1.44kVA.
The proposed topology introduces two more slow diodes
(Da and Db) to the bridgeless configuration to link the
ground of the PFC to the input line. However, the current
does not always return through these diodes, so their
associated conduction losses are low. This occurs since the
inductors exhibit low impedance at the line frequency, a
large portion of the current flows through the FET intrinsic
body diodes. Also the gating signals for FETs are 180° out of
phase.
A detailed converter description and steady-state
operation analysis is given in the following section.
Figure 4. Phase shifted semi-bridgeless PFC boost topology
III.
OPERATING PRINCIPLE AND STEADY-STATE
ANALYSIS
To analyze the circuit operation, the input line cycle has
been separated into the positive and negative half-cycles as
explained in sub-sections A and B that follow. In addition,
the detailed circuit operation depends on the duty cycle.
Positive half-cycle operation analysis is provided for D > 0.5
in sub-section C and D < 0.5 in sub-section D.
A. Positive Half-Cycle Operation
Referring to Fig. 4, during the positive half-cycle, when
the AC input voltage is positive, Q1 turns on and current
flows through L1 and Q1 and continues through Q2 and then
L2, returning to the line while storing energy in L1 and L2.
When Q1 turns off, energy stored in L1 and L2 is released as
current flows through D1, through the load and returns
through the body diode of Q2/partially through Db back to
the input.

3. B. Negative Half-Cycle Operation
Referring to Fig. 4, during the negative half-cycle, when
the AC input voltage is negative, Q2 turns on and current
flows through L2 and Q2 and continues through Q1 and then
L1, returning to the line while storing energy in L2 and L1.
When Q2 turns off, energy stored in L2 and L1 is released as
current flows through D2, through the load and returns split
between the body diode of Q1 and Da back to the input.
the ripple current components are derived, enabling
calculation of the input ripple current, which provides design
guidance to meet the required input current ripple standard.
C. Detailed Positive Half-Cycle Operation and Analysis for
D > 0.5
The detailed operation of the proposed converter depends
on the duty cycle. During any half-cycle, the converter duty
cycle is either greater than 0.5 (when the input voltage is
smaller than half of output voltage) or smaller than 0.5
(when the input voltage is greater than half of output
voltage). The three unique operating interval circuits of the
proposed converter are provided in Fig. 5 to Fig. 7 for duty
cycles larger than 0.5 during the positive half-cycle.
Figure 5. Interval 1and 3: Q1 and Q2 are ON
Figure 8. Phase shifted semi-bridgeless boost converter steady-state
Waveforms at D > 0.5
Figure 6. Interval 2: Q1 ON, body diode of Q2 conducting
Interval 1 [t0-t1]: At t0, Q1/ Q2 are on, as shown in Fig.5.
During this interval, the current in series inductances L1 and
L2 increases linearly and stores the energy in these inductors.
The energy stored in Co provides energy to the load. The
ripple currents in Q1 and Q2 are the same as the current in
series inductances L1 and L2, where the ripple current is
given by:
∆I
Figure 7. Interval 4: Q1 OFF and Q2 ON
Waveforms of the proposed converter during positive
half-cycle operation with D>0.5 are shown in Fig. 8. The
intervals of operation are explained as follows. In addition,
L
L
v D
T
(1)
Interval 2 [t1-t2]: At t1, Q1 is on and Q2 is off, as shown
in Fig.6. During this interval, the current in series
inductances L1 and L2 continues to increase linearly and
store the energy in these inductors. The energy stored in Co
provides the load energy. The ripple currents in Q1 and body
diode of Q2 are the same as the current in series inductances
L1 and L2, where the ripple current is given by:
∆I
L
L
v 1
D T
(2)

4. Interval 3 [t2-t3]: At t2, Q1/Q2 are on again, and interval
1 is repeated, as shown in Fig. 5. During this interval, the
current in series inductances L1 and L2 increases linearly
and stores the energy in these inductors. The ripple currents
in Q1 and Q2 are the same as the ripple current in series
inductances L1 and L2, as shown in equation (1).
are shown in Fig. 12. The intervals of operation are
explained as follows.
Interval 4 [t3-t4]: At t3, Q1 is off and Q2 is on, as shown I
Fig. 7. During this interval, the energy stored in L1 and L2 is
released to the output through L1, D1, Q2 and L2. The ripple
currents in D1 and Q2 are the same as the ripple currents in
L1 and L2:
∆
1
(3)
Figure 9. Interval 1and 3: Q1 and Q2 are OFF, body diode of Q2
conducting
Figure 12. Phase shifted semi-bridgeless boost converter steady-state
waveforms at D < 0.5
Figure 10. Interval 2: Q1 ON, body diode of Q2 conducting
Interval 1 [t0-t1]: At t0, Q1/ Q2 are off, as shown in Fig.9.
During this interval, the energy stored in L1 and L2 are
released to the output through L1, D1, body diode of Q2 and
L2. The ripple currents in D1 and body diode of Q2 are the
same as the ripple currents in L1 and L2:
∆I
Figure 11. Interval 4: Q1 OFF and Q2 ON
D. Detailed Positive Half-Cycle Operation and Analysis for
D < 0.5
The three unique operating interval circuits of the
proposed converter are given in Fig. 9 to Fig. 11 for duty
cycles smaller than 0.5 during the positive half-cycle. The
waveforms of the proposed converter during these conditions
L
D T
L
(4)
Interval 2 [t1-t2]: At t1, Q1 is on and Q2 is off, as shown
in Fig.10. During this interval, the current in series
inductances L1 and L2 continues to increase linearly and
store the energy in these inductors. The energy stored in Co
provides energy to the load. The ripple currents in Q1 and
the body diode of Q2 are the same as the current in series
inductances L1 and L2, where the ripple current is given by:
∆I
L
L
v DT
(5)
Interval 3 [t2-t3]: At t2, Q1/Q2 are off again, and interval
1 is repeated, as shown in Fig. 9. During this interval, the
current in series inductances L1 and L2 increases linearly

5. (6)
∆
The operation of converter during the negative input
voltage half-cycle is similar to the operation of converter
during the positive input voltage half-cycle.
IV.
LOSS EVALUATION
The estimated loss distribution of the semiconductors is
provided in Fig. 13 at 70 kHz switching frequency, 240V
input and 3300W load for benchmark conventional boost and
interleaved boost converters and the proposed phase shifted
semi-bridgeless boost converter. The currents in regular
diodes Da and Db were assumed to be split with the current
going through intrinsic body diodes for phase shifted semibridgeless topology. The regular diodes in input bridge
rectifiers have the largest share of losses among the
topologies with the input bridge rectifier. The phase shifted
semi-bridgeless topology nearly eliminates this large loss
component (~30W). However, the tradeoff is that the FET
losses are higher and the intrinsic body diodes of FETs
conduct, producing new losses (~8W). The fast diodes in the
conventional and interleaved PFC have slightly lower power
losses, since the boost RMS current is higher in these
topologies.
47.7W
42.3W
39.4
60
Conventional Boost
Interleaved Boost
TABLE I. DEVICES/COMPONENTS USED IN EXPERIMENTAL PROTOTYPES
Topology
Components Used in Prototype Unit Head
Device
Part # / Value
# of Devices
25ETS08S
2
Fast Diode
IDB06S60C
2
MOSFET
IPB60R099CP
2
Inductors
400 μH
2
25ETS08S
4
Fast Diode
IDB06S60C
2
MOSFET
IPB60R099CP
2
Inductors
400 μH
2
Regular Diode
Regular Diode
Pictures of the proposed phase shifted bridgeless boost
prototype are provided in Fig. 14. It consists of a control
board, a capacitor bank of 820 μF and an IMS power board
attached to a heatsink with the PFC inductors.
Phase Shifted Semi-Bridgeless Boost
6.9W
6.9W
9.8
4.1
20
10.8W
5.4W
21.6
30
10
Prototypes of a phase shifted bridgeless boost converter
and an interleaved boost converter were built to verify the
proof-of-concept and analytical work presented in this paper
and to benchmark the proposed converter. The devices used
in experimental prototypes are provided in Table 1.
0.0W
0.0W
4.1
40
30.0W
30.0W
Power Losses (W)
50
EXPERIMENTAL RESULTS
V.
Phase Shifted
Semi-bridgeless
PFC converter
Interval 4 [t3-t4]: At t3, Q1 is off and Q2 is on, as shown I
Fig. 11. During this interval, the energy stored in L1 and L2
is released to the output through L1, D1, Q2 and L2. The
ripple currents in D1 and Q2 are the same as the ripple
currents in L1 and L2:
proposed phase shifted semi-bridgeless boost are 17% lower
than the benchmark conventional boost and 7% lower than
the interleaved boost . Since the benchmark converter bridge
rectifier losses are large, it is expected that phase shifted
semi-bridgeless boost converter should have the lowest
losses among the topologies investigated. Additionally, it is
noted that the losses in the input bridge rectifiers are 63% of
total losses in the conventional PFC converter and 71% of
total losses in the interleaved PFC converter. Therefore,
eliminating the input bridge in PFC converters is justified
despite that the introduction of new losses.
Interleaved PFC
converter
and stores the energy in these inductors. The ripple currents
in D1 and body diode of Q2 are the same as the ripple
current in series inductances L1 and L2, as shown in
equation (1).
Total
Losses
Intrinsic
Body
Diodes
FETs
Fast
Diodes
Regular
Diodes
0
Semiconductor Losses
Figure 13. Comparison of the estimated loss distribution in the
semiconductors at 70kHz switching frequency, 240V input, 3300W load at
400V
Overall the FETs are under slightly more stress in phase
shifted semi-bridgeless topology, but the total loss for the
Figure 14. Top: control board, Bottom: power board

6. 100
98
40
30
20
10
0
3500
Output Power (W)
Figure 17. Efficiency as a function of output power at Vin = 120V,
Vo=400V and 70kHz switching frequency
3500
Vin=120
3000
1800
1600
Output Power (W)
1400
1200
1000
800
600
400
200
0
91
Vin=240
2500
Phase Shifted SemiBridgeless PFC Converter
2000
Interleaved PFC Converter
1500
95
0
Power factor
96
1.02
1
0.98
0.96
0.94
0.92
0.9
0.88
0.86
0.84
1000
97
92
1800
Figure 19. THD as a function of output power at Vin = 120 V and 240V,
Vo=400V and 70kHz switching frequency
98
93
1600
Output Power (W)
Figure 16. Loss reduction as a function of output power at Vin = 240V,
Vo=400V and 70kHz switching frequency
500
3000
2500
2000
1500
1000
500
0
0
3500
10
Vin=120
3000
20
Vin=240
2500
THD (%)
30
2000
40
45
40
35
30
25
20
15
10
5
0
1500
Loss Reduction for PFC
Converters at Vin = 240 V
1000
70
From the results, it is noted that proposed semi-bridgeless
PFC converter achieves a peak efficiency of 98.6% at 1 kW
output power. Additionally, the light load efficiency of the
proposed converter is significantly better than that of the
interleaved PFC due to the absence of input bridge rectifier.
However, as the load increases, the efficiency drops due to
additional heat dissipation in the intrinsic body diodes of the
FETs.
500
3500
3000
2500
2000
1500
1000
500
0
Figure 15. Efficiency as a function of output power at Vin = 240V,
Vo=400V and 70kHz switching frequency
94
1400
Figure 18. Loss reduction as a function of output power at Vin = 120V,
Vo=400V and 70kHz switching frequency
Output Power (W)
50
1200
Output Power (W)
Phase Shifted SemiBridgeless PFC Converter
60
1000
Interleaved PFC Converter
96
800
600
400
200
0
94
Loss Reduction (%)
Loss Reduction for PFC
Converters at Vin = 120 V
50
97
95
Efficiency (%)
60
0
Efficiency (%)
99
70
Loss Reduction (%)
The experimental efficiency of the phase shifted
bridgeless boost converter and benchmark interleaved boost
converter is provided in Fig. 15 for 240V input and Fig. 17
for 120V input at 70 kHz switching frequency and 400 V
output. Loss reduction curves as a function of output power
are provided in Fig. 16 and Fig. 18 for 240V and 120V input,
respectively.
Output Power (W)
Figure 20. Power Factor as a function of output power at Vin = 120 V and
240V, Vo=400V and 70kHz switching frequency

7. 2.5
Amplitude (A)
2
EN 61000-3-2 Class D Limits (A)
Output Voltage
1.5
Amplitude (A) Vin = 120 V
1
Amplitude (A) Vin = 240 V
0.5
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
35
37
39
0
Harmonics Order
Inductor
Current
Input Voltage
Input Current
Figure 21. Harmonics orders at Vin = 120 V and 240V, compared against
EN61000-3-2 standard.
In order to verify the quality of the input current, the input
current THD is shown in Fig.19. The power factor and
harmonic orders are given and compared with EN 61000-3-2
standard in Fig.20 and 21. It is noted that mains current THD
is less than 5% from 50% load to full load and it is compliant
to IEC 6100-3-2 (Fig. 19 and Fig. 21). The converter power
factor is shown over entire load range for 120 and 240V
input in Fig. 20. The power factor is greater than 0.99 from
50% load to full load.
Experimental waveforms from the proposed converter
prototype are provided in Fig. 22 through Fig. 26. The input
current, input voltage and output voltage are given in Fig. 22.
As it can be seen, the input current is in phase with the input
voltage and has a sinusoidal shape. Additionally, there is a
low frequency ripple on output voltage, which is inversely
proportional to the value of PFC bus output capacitors.
In Fig. 23, the inductor current is provided in addition to
the above mentioned waveforms from Fig. 22. It is noted that
during the positive half-cycle, the inductor current is the
same as input current. However, during the negative halfcycle, the input current is partially flowing through slow
diodes, Da and Db.
Figure 23. Inut current, inducotr current, input voltage and output voltage.
Ch1= Vo 100V/div. Ch2= Vin 100V/div. Ch3= IL1 10A/div. Ch4= Iin
10A/div.
Sensed FET Current
Input
Current
Inductor
Current
Figure 24. Inductor current, input current and sensed FET current.
Ch1= Sensed IQ1 2V/div. Ch3= IL1 / IDb 10A/div. Ch4= Iin 10A/div
Inductor Current
Output Voltage
Sensed FET
Current
Input Voltage
Gating Signal
Input Current
Figure 22. Inut current, input voltage and output voltage.
Ch1= Vo 100V/div. Ch2= Vin 100V/div. Ch4= Iin 10A/div.
Figure 25. Gating signal, Inductor and sensed FET current for D < 0.5
Ch1= Vg 10V/div. Ch2= IQ1 2V/div. Ch3= IL1 10A/div
In Fig.24 the inductor current, input current and current
sensed in the FET through a current transformer are given.

8. The gating signals, sensed FET current and the inductor
current are provided for duty cycles less than 0.5, Fig 25, and
greater than 0.5, Fig. 26. These waveforms match the
theoretical models.
[2]
[3]
[4]
Inductor Current
[5]
Sensed FET Current
[6]
Gating Signal
Figure 26. Gating signal, Inductor and sensed FET current for D > 0.5
Ch1= Vg 10V/div. Ch2= IQ1 2V/div. Ch3= IL1 10A/div
VI.
CONCLUSIONS
A new high performance phase shifted semi-bridgeless
AC-DC Boost converter topology has been presented in this
paper for the front-end AC-DC converter in PHEV battery
chargers. The proposed converter features high efficiency at
light loads and low lines, which is critical to minimize the
charger size, charging time and the amount and cost of
electricity drawn from the utility; the component count,
which reduces the charger cost; and reduced EMI. The
converter is ideally suited for automotive level I residential
charging applications in North America where the typical
supply is limited to 120V and 1.44kVA.
An analysis and performance characteristics are presented.
A breadboard converter circuit has been built to verify the
proof-of-concept. The theoretical waveforms were compared
with the results taken from prototype unit. Additionally, key
experimental waveforms were provided and input current
harmonics at each harmonic order were compared more
explicitly with the IEC 6100-3-2 standard limits.
Experimental results demonstrate that the mains current
THD is smaller than 5% from 50% load to full load and the
converter is compliant with the IEC 6100-3-2 standard. The
converter power factor was also provided for full power
range at 120 and 240V input. The power factor is greater
than 0.99 from 50% load to full load. The proposed
converter achieves a peak efficiency of 98.6 % at 240 V
input and 1 kW output power.
REFERENCES
[1]
Young-Joo Lee; Khaligh, A.; Emadi, A.; "Advanced Integrated
Bidirectional AC-DC and DC-DC Converter for Plug-In Hybrid
Electric Vehicles," IEEE Trans. on Vehicular Technology, vol. 58,
pp. 3970 - 3980 2009.
[7]
[8]
[9]
[10]
[11]
[12]
[13]
[14]
[15]
[16]
[17]
[18]
[19]
K. Morrow, D. Karner, and J. Francfort, "Plug-in Hybrid Electric
Vehicle Charging Infrastructure Review," U.S. Departent of Energy Vehicle Technologies Program, 2008.
Petersen, L.; Andersen, M.; "Two-Stage Power Factor Corrected
Power Supplies: The Low Component-Stress Approach " in Proc.
IEEE Applied Power Electronics Conference and Exposition, APEC.
vol. 2, 2002, pp. 1195 - 1201.
Singh, B.; Singh, B.N.; Chandra, A.; Al-Haddad, K.; Pandey, A.;
Kothari, D.P.; "A Review of Single-Phase Improved Power Quality
AC-DC Converters," IEEE Trans. on Industrial Electronics, vol. 50,
pp. 962 - 981, 2003.
Dehong Xu; Jindong Zhang; Weiyun Chen; Jinjun Lin; Lee, F.C.;
"Evaluation of output filter capacitor current ripples in single phase
PFC converters " in Proc. IEEE Power Conversion Conference, PCC.
vol. 3 Osaka, Japan, 2002, pp. 1226 - 1231.
Lu, B.; Brown, R.; Soldano, M.; "Bridgeless PFC implementation
using one cycle control technique," in Proc. IEEE Applied Power
Electronics Conference and Exposition, APEC, vol. 2, 2005, pp. 812 817.
Petrea, C.; Lucanu, M.; "Bridgeless Power Factor Correction
Converter Working at High Load Variations," in Proc. International
Symposium on Signals, Circuits and Systems, ISSCS. vol. 2, 2007, pp.
1-4
U. Moriconi, "A Bridgeless PFC Configuration based on L4981 PFC
Controller ": STMicroelectronics Application Note AN1606, 2002.
J. M. Hancock, "Bridgeless PFC Boosts Low-Line Efficiency,"
Infineon Technologies, 2008.
Yungtaek Jang; Jovanovic, M.M.; Dillman, D.L.; "Bridgeless PFC
boost rectifier with optimized magnetic utilization," in Proc. IEEE
Applied Power Electronics Conference and Exposition, APEC, 2008,
pp. 1017 – 1021.
Yungtaek Jang; Jovanovic, M.M.; "A Bridgeless PFC Boost Rectifier
With Optimized Magnetic Utilization," IEEE Trans. on Power
Electronics, vol. 24, pp. 85 - 93 2009.
Woo-Young Choi; Jung-Min Kwon; Eung-Ho Kim; Jong-Jae Lee;
Bong-Hwan Kwon; "Bridgeless Boost Rectifier With Low
Conduction Losses and Reduced Diode Reverse-Recovery Problems,"
IEEE Trans. on Industrial Electronics, vol. 54, pp. 769 – 780, April
2007.
Huber, L.; Yungtaek Jang; Jovanovic, M.M.; "Performance
Evaluation of Bridgeless PFC Boost Rectifiers," IEEE Trans. on
Power Electronics, vol. 23, pp. 1381 - 1390 2008.
Pengju Kong; Shuo Wang; Lee, F.C.; "Common Mode EMI Noise
Suppression for Bridgeless PFC Converters," IEEE Trans. on Power
Electronics, vol. 23, pp. 291 – 297, January 2008 2008.
Baur, T.; Reddig, M.; Schlenk, M.; "Line-conducted EMI-behaviour
of a High Efficient PFC-stage without input rectification," Infineon
Technology – Application Note, 2006.
Frank, W.; Reddig, M.; Schlenk, M.; "New control methods for
rectifier-less PFC-stages," in Proc. EEE International Symposium on
Industrial Electronics. vol. 2, 2005, pp. 489 - 493
M. O’Loughlin;, "An Interleaved PFC Preregulator for High-Power
Converters." vol. Topic 5: Texas Instrument Power Supply Design
Seminar, 2007, pp. 5-1, 5-14.
Yungtaek Jang; Jovanovic, M.M.; "Interleaved Boost Converter With
Intrinsic Voltage-Doubler Characteristic for Universal-Line PFC
Front End," IEEE Trans. on Power Electronics, vol. 22, pp. 1394 –
1401, July 2007.
Balogh, L.; Redl, R.; "Power-factor correction with interleaved boost
converters in continuous-inductor-current mode," in Proc. IEEE
Applied Power Electronics Conference and Exposition, 1993, pp. 168
- 174.